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TM
MP2106
1.5A, 15V, 800KHz Synchronous Buck Converter
The Future of Analog IC Technology
TM
DESCRIPTION
The MP2106 is a 1.5A, 800KHz synchronous buck converter designed for low voltage applications requiring high efficiency. It is capable of providing output voltages as low as 0.9V, and integrates top and bottom switches to minimize power loss and component count. The 800KHz switching frequency reduces the size of filtering components, further reducing the solution size. The MP2106 includes cycle-by-cycle current limiting and under voltage lockout. The internal power switches, combined with the tiny 10-pin MSOP and QFN packages, provide a solution requiring a minimum of surface area.
FEATURES
* * * * * * * * * * * * * * * * * 1.5A Output Current Synchronous Rectification Internal 210m and 255m Power Switches Input Range of 2.6V to 15V >90% Efficiency Zero Current Shutdown Mode Under Voltage Lockout Protection Soft-Start Operation Thermal Shutdown Internal Current Limit (Source & Sink) Tiny 10-Pin MSOP or QFN Package DC/DC Regulation from Wall Adapters Portable Entertainment Systems Set Top Boxes Digital Video Cameras, DECT Networking Equipment Wireless Modems
APPLICATIONS
EVALUATION BOARD REFERENCE
Board Number EV2106DQ/DK-00A Dimensions 2.5"X x 2.0"Y x 0.5"Z
"MPS" and "The Future of Analog IC Technology" are Trademarks of Monolithic Power Systems, Inc.
TYPICAL APPLICATION
INPUT 2.6V to 15V R4 OFF ON
5 1 3
C7 C1
7 VIN RUN SS COMP VREF 4 10 C6 10nF 6 BST LX 8 2
Efficiency vs. Load Current
100 90 80
EFFICIENCY (%)
10nF
VIN=3.3V
L1 OUTPUT 1.8V / 1.5A
70 60 50 40 30 20 10 0 0.01 0.1 1 LOAD CURRENT (A) 10 VIN=5V
MP2106
FB SGND PGND 9
C5 10nF
C3 3.3nF R3
R2 R1
C2
MP2106_TAC_S01
MP2106_TAC_EC02
MP2106 Rev. 1.6 2/22/2006
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TM
MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER
PACKAGE REFERENCE
TOP VIEW
SS FB COMP VREF RUN 1 2 3 4 5 10 9 8 7 6 SGND PGND LX VIN BST
TOP VIEW
SS FB COMP VREF RUN 1 2 3 4 5 10 9 8 7 6 SGND PGND LX VIN BST
MP2106_PD01-MSOP10
EXPOSED PAD ON BACKSIDE
MP2106_PD02-QFN10
Part Number* MP2106DK *
Package MSOP10
Temperature -40C to +85C
Part Number** MP2106DQ
Package QFN10 (3mm x 3mm)
Temperature -40C to +85C
For Tape & Reel, add suffix -Z (eg. MP2106DK-Z) For Lead Free, add suffix -LF (eg. MP2106DK-LF-Z)
** For Tape & Reel, add suffix -Z (eg. MP2106DQ-Z)
For Lead Free, add suffix -LF (eg. MP2106DQ-LF-Z)
ABSOLUTE MAXIMUM RATINGS (1)
Input Supply Voltage VIN .............................. 16V LX Voltage VLX ..................... -0.3V to VIN + 0.3V BST to LX Voltage ......................... -0.3V to +6V Voltage on All Other Pins............... -0.3V to +6V Storage Temperature...............-55C to +150C
Thermal Resistance
(3)
MSOP10 ................................ 150 ..... 65... C/W QFN10 .................................... 50 ...... 12... C/W
Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1" square of 1 oz copper.
JA
JC
Recommended Operating Conditions
(2)
Input Supply Voltage VIN ..................2.6V to 15V Output Voltage VOUT ........................0.9V to 5.5V Operating Temperature..............-40C to +85C
ELECTRICAL CHARACTERISTICS
VIN = 5.0V, TA = +25C, unless otherwise noted.
Parameter Input Voltage Range Input Under Voltage Lockout Input Under Voltage Lockout Hysteresis Shutdown Supply Current Operating Supply Current VREF Voltage RUN Input Low Voltage RUN Input High Voltage RUN Hysteresis RUN Input Bias Current Oscillator Switching Frequency Maximum Duty Cycle Minimum On Time
MP2106 Rev. 1.6 2/22/2006
Symbol Condition VIN
Min 2.6
Typ 2.2 100
Max 15
Units V V mV
VREF VIL VHL
VRUN 0.3V VRUN > 2V, VFB = 1.1V VIN = 2.6V to 15V 1.5
0.5 1.2 2.4
1.0 1.8 0.4
100 1 fSW DMAX tON 700 85 800 200 900
A mA V V V mV A KHz % ns 2
VFB = 0.7V
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TM
MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 5.0V, TA = +25C, unless otherwise noted.
Parameter Error Amplifier Voltage Gain Transconductance COMP Maximum Output Current FB Regulation Voltage FB Input Bias Current Soft-Start Soft-Start Current Output Switch On-Resistance Switch On Resistance Synchronous Rectifier On Resistance Switch Current Limit (Source) Synchronous Rectifier Current Limit (Sink) Thermal Shutdown Symbol Condition AVEA GEA VFB IFB ISS VIN = 5V VIN = 3V VIN = 5V VIN = 3V 875 VFB = 0.895V Min Typ 400 300 30 895 -100 2 255 315 210 255 2.5 350 160 Max Units V/V A/V A mV nA A m m m m A mA C
915
PIN FUNCTIONS
Pin # 1 2 3 4 5 6 7 8 9 10 Name Description Soft-Start Input. Place a capacitor from SS to SGND to set the soft-start period. The MP2106 SS sources 2A from SS to the soft-start capacitor at startup. As the SS voltage rises, the feedback threshold voltage increases to limit inrush current during startup. Feedback Input. FB is the inverting input of the internal error amplifier. Connect a resistive FB voltage divider from the output voltage to FB to set the output voltage value. Compensation Node. COMP is the output of the error amplifier. Connect a series RC network COMP to compensate the regulation control loop. Internal 2.4V Regulator Bypass. Connect a 10nF capacitor between VREF and SGND to VREF bypass the internal regulator. Do not apply any load to VREF. On/Off Control Input. Drive RUN high to turn on the MP2106; low to turn it off. For automatic RUN startup, connect RUN to VIN via a pullup resistor. Power Switch Boost. BST powers the gate of the high-side N-Channel power MOSFET switch. BST Connect a 10nF or greater capacitor between BST and LX. Internal Power Input. VIN supplies the power to the MP2106 through the internal LDO VIN regulator. Bypass VIN to PGND with a 10F or greater capacitor. Connect VIN to the input source voltage. Output Switching Node. LX is the source of the high-side N-Channel switch and the drain of the LX low-side N-Channel switch. Connect the output LC filter between LX and the output. Power Ground. PGND is the source of the N-Channel MOSFET synchronous rectifier. Connect PGND PGND to SGND as close to the MP2106 as possible. SGND Signal Ground.
MP2106 Rev. 1.6 2/22/2006
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TM
MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
Circuit of Figure 2, VIN = 5V, VOUT = 1.8V, L1 = 5H, C1 = 10F, C2 = 22F, TA = +25C, unless otherwise noted.
Steady State Operation
1.5A Load
VSW 5V/div. VO AC Coupled 10mV/div. VIN AC Coupled 200mV/div. VSW 5V/div. VO AC Coupled 10mV/div. VIN AC Coupled 20mV/div. IL 1A/div.
Steady State Operation
No Load
Load Transient
VOUT AC Coupled 200mV/div.
IL 1A/div.
ILOAD 1A/div.
IL 1A/div.
MP2106-TPC01
MP2106-TPC02
MP2106-TPC03
Startup from Shutdown
1.5A Resistive Load
VEN 2V/div. VOUT 1V/div. IL 1A/div. VSW 5V/div. VEN 2V/div. VOUT 1V/div. IL 1A/div. VSW 5V/div.
Startup from Shutdown
No Load
1ms/div.
MP2106-TPC04
1ms/div.
MP2106-TPC05
Short Circuit Protection
Short Circuit Recovery
VOUT 1V/div.
VOUT 1V/div.
IL 1A/div.
IL 1A/div.
MP2106-TPC06
MP2106-TPC07
MP2106 Rev. 1.6 2/22/2006
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TM
MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER
OPERATION
VIN VIN 2.6V to 15V
7
C1
OFF ON
RUN
5
ENABLE CKT & LDO REGULATOR
GATE DRIVE REGULATOR
Vdr
CURRENT SENSE AMPLIFIER
+ --
VREF C6
4
VBP 2.4V
PWM COMPARATOR + --
Vdr
6
BST C7 LX Vdr
8
L1 VOUT C2
CONTROL LOGIC
800KHz OSCILLATOR RAMP VBP CURRENT LIMIT COMPARATOR UVLO & THERMAL SHUTDOWN + -+ -9
R2 PGND
SS C5
1
-GM -ERROR AMPLIFIER +
2
FB
CURRENT LIMIT THRESHOLD
10
VFB 0.895V R1
3
SGND C4
COMP R3 C3
MP2106_BD01
Figure 1--Functional Block Diagram
MP2106 Rev. 1.6 2/22/2006
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MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER The MP2106 measures the output voltage through an external resistive voltage divider and compares that voltage to the internal 0.9V reference in order to generate the error voltage at COMP. The current-mode regulator uses the voltage at COMP and compares it to the inductor current to regulate the output voltage. The use of current-mode regulation improves transient response and improves control loop stability. At the beginning of each cycle, the high-side N-Channel MOSFET is turned on, forcing the inductor current to rise. The current at the drain of the high-side MOSFET is internally measured and converted to a voltage by the current sense amplifier. That voltage is compared to the error voltage at COMP. When the inductor current rises sufficiently, the PWM comparator turns off the high-side switch and turns on the low-side switch, forcing the inductor current to decrease. The average inductor current is controlled by the voltage at COMP, which in turn is controlled by the output voltage. Thus the output voltage controls the inductor current to satisfy the load. Since the high-side N-Channel MOSFET requires voltages above VIN to drive its gate, a bootstrap capacitor from LX to BST is required to drive the high-side MOSFET gate. When LX is driven low (through the low-side MOSFET), the BST capacitor is internally charged. The voltage at BST is applied to the high-side MOSFET gate to turn it on, and maintains that voltage until the high-side MOSFET is turned off and the low-side MOSFET is turned on, and the cycle repeats. Connect a 10nF or greater capacitor from BST to SW to drive the high-side MOSFET gate.
APPLICATION INFORMATION
INPUT 2.6V to 15V
5 1 3
C7 10nF
7 VIN RUN SS COMP VREF 4 10 6 BST LX 8 2
MP2106
FB SGND PGND 9
OUTPUT 1.8V / 1.5A
C5 10nF
C4 OPEN
C3 3.3nF
C6 10nF
MP2106_TAC_F02
Figure 2--Typical Application Circuit Internal Low-Dropout Regulator The internal power to the MP2106 is supplied from the input voltage (VIN) through an internal 2.4V low-dropout linear regulator, whose output is VREF. Bypass VREF to SGND with a 10nF or greater capacitor for proper operation. The internal regulator can not supply more current than is required to operate the MP2106. Therefore, do not apply any external load to VREF. Soft-Start The MP2106 includes a soft-start timer that slowly ramps the output voltage at startup to prevent excessive current at the input. When power is applied to the MP2106, and RUN is asserted, a 2A internal current source charges the external capacitor at SS. As the capacitor charges, the voltage at SS rises. The MP2106 internally limits the feedback threshold voltage at FB to that of the voltage at SS. This forces the output voltage to rise at the same rate as the voltage at SS, forcing the output
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MP2106 Rev. 1.6 2/22/2006
www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. (c) 2006 MPS. All Rights Reserved.
TM
MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER voltage to ramp linearly from 0V to the desired regulation voltage during soft-start. The soft-start period is determined by the equation:
t SS = 0.45 x C5
capacitors should be placed close to the MP2106. For most applications, a 10F ceramic capacitor will work. Selecting the Output Capacitor The output capacitor (C2) is required to maintain the DC output voltage. Low ESR capacitors are preferred to keep the output voltage ripple to a minimum. The characteristics of the output capacitor also affect the stability of the regulation control system. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. The output voltage ripple is:
VRIPPLE = VOUT V x 1 - OUT f SW x L VIN 1 x R ESR + 8 x f SW x C2
Where C5 (in nF) is the soft-start capacitor from SS to GND, and tSS (in ms) is the soft-start period. Determine the capacitor required for a given soft-start period by the equation:
C5 = 2.22 x t SS
Use values between 10nF and 22nF for C5 to set the soft-start period (between 4ms and 10ms). Setting the Output Voltage (see Figure 2) Set the output voltage by selecting the resistive voltage divider ratio. The voltage divider drops the output voltage to the 0.895V feedback voltage. Use 10k for the low-side resistor of the voltage divider. Determine the high-side resistor by the equation:
V R2 = OUT - 1 x R1 0.895 V
Where VRIPPLE is the output voltage ripple, fSW is the switching frequency, VIN is the input voltage, RESR is the equivalent series resistance of the output capacitors and fSW is the switching frequency. Choose an output capacitor to satisfy the output ripple requirements of the design. A 22F ceramic capacitor is suitable for most applications. Selecting the Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor results in less ripple current that will result in lower output ripple voltage. However, the larger value inductor is likely to have a larger physical size and higher series resistance. Choose an inductor that does not saturate under the worst-case load conditions. A good rule for determining the inductance is to allow the peak-to-peak ripple current to be approximately 30% to 40% of the maximum load current. Make sure that the peak inductor current (the load current plus half the peak-topeak inductor ripple current) is below 2.5A to prevent loss of regulation due to the current limit.
Where R2 is the high-side resistor, VOUT is the output voltage and R1 is the low-side resistor. Selecting the Input Capacitor The input current to the step-down converter is discontinuous, and so a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. A low ESR capacitor is required to keep the noise at the IC to a minimum. Ceramic capacitors are preferred, but tantalum or low ESR electrolytic capacitors may also suffice. The capacitor can be electrolytic, tantalum or ceramic. Because it absorbs the input switching current it must have an adequate ripple current rating. Use a capacitor with RMS current rating greater than 1/2 of the DC load current. For stable operation, place the input capacitor as close to the IC as possible. A smaller high quality 0.1F ceramic capacitor may be placed closer to the IC with the larger capacitor placed further away. If using this technique, it is recommended that the larger capacitor be a tantalum or electrolytic type. All ceramic
MP2106 Rev. 1.6 2/22/2006
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MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER Calculate the required inductance value by the equation:
L= VOUT x (VIN - VOUT ) VIN x f SW x I
The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). The zero is:
f Z1 = 1 2 x R3 x C3
Where I is the peak-to-peak inductor ripple current. It is recommended to choose I to be 30%~40% of the maximum load current. Compensation The system stability is controlled through the COMP pin. COMP is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC loop gain is:
V A VDC = FB V OUT x A VEA x G CS x R LOAD
If large value capacitors with relatively high equivalent-series-resistance (ESR) are used, the zero due to the capacitance and ESR of the output capacitor can be compensated by a third pole set by R3 and C4. The pole is:
fP3 = 1 2 x R3 x C4
Where VFB is the feedback voltage, AVEA is the transconductance error amplifier voltage gain, GCS is the current sense transconductance (roughly the output current divided by the voltage at COMP) and RLOAD is the load resistance:
R LOAD V = OUT I OUT
The system crossover frequency (the frequency where the loop gain drops to 1, or 0dB, is important. Set the crossover frequency to below one tenth of the switching frequency to insure stable operation. Lower crossover frequencies result in slower response and worse transient load recovery. Higher crossover frequencies degrade the phase and/or gain margins and can result in instability. Table 1--Compensation Values for Typical Output Voltage/Capacitor Combinations
VOUT C2 R3 C3 C4
Where IOUT is the output load current. The system has 2 poles of importance, one is due to the compensation capacitor (C3), and the other is due to the load resistance and the output capacitor (C2), where:
fP1 = GEA 2 x A VEA x C3
1.8V 22F Ceramic 2.5V 22F Ceramic 3.3V 22F Ceramic 47F Tantalum 1.8V (300m) 47F Tantalum 2.5V (300m) 47F Tantalum 3.3V (300m)
6.8k 9.1k 12k 13k 18k 24k
3.3nF 2.2nF 1.8nF 2nF 1.2nF 1nF
None None None 1nF 750pF 560pF
P1 is the first pole, and GEA is the error amplifier transconductance (300A/V) and
fP 2 = 1 2 x R LOAD x C2
Choosing the Compensation Components The values of the compensation components given in Table 1 yield a stable control loop for the given output voltage and capacitor. To optimize the compensation components for conditions not listed in Table 1, use the following procedure.
MP2106 Rev. 1.6 2/22/2006
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MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER Choose the compensation resistor to set the desired crossover frequency. Determine the value by the following equation:
R3 = 2 x C2 x VOUT x f C G EA x G CS x VFB
External Boost Diode For input voltages less than or equal to 5V, it is recommended that an external boost diode be added. This will help improve the regulator efficiency. The diode can be a low cost diode such as an IN4148 or BAT54.
5V BOOST DIODE 10nF 8
MP2106_F03
Where fC is the desired crossover frequency (preferably 33KHz). Choose the compensation capacitor to set the zero below one fourth of the crossover frequency. Determine the value by the following equation:
2 C3 > x R3 x f C
BST
6
MP2106
LX
Figure 3--External Boost Diode This diode is also recommended for high duty cycle operation (when
VOUT >65%) and high VIN
Determine if the second compensation capacitor, C4 is required. It is required if the ESR zero of the output capacitor happens at less than half of the switching frequency. Or:
x C2 x R ESR x f SW > 1
If this is the case, then add the second compensation capacitor. Determine the value by the equation:
C4 = C2 x R ESR(max) R3
applications. output voltage (VOUT>12V) However, do not exceed the absolute maximum voltage for these pins.
Where RESR(MAX) is the maximum ESR of the output capacitor.
MP2106 Rev. 1.6 2/22/2006
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MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER
PACKAGE INFORMATION
MSOP10
MP2106 Rev. 1.6 2/22/2006
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MP2106 - 1.5A, 15V, 800KHz SYNCHRONOUS BUCK CONVERTER
QFN10 (3mm x 3mm)
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications.
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